1. Field of the Invention
The field of this invention relates to receiver front end circuits comprising (at least) low-noise amplifiers and down-conversion mixers. The invention is applicable to, but not limited to, receiver front end circuits comprising inductor-less low-noise amplifiers and noise cancellation circuits for wireless communication units.
2. Background of the Invention
In the field of wireless communications, a communication receiver is a key component of a communications unit that is designed to receive and process wirelessly received signals. Often the signals are received at very low power, due to signal propagation losses attributed to transmit signals that are wirelessly radiated from a distal wireless transmitter. Thus, by the time that the wirelessly transmitted signal has reached the receiver, the amount of received signal power is very low.
Once acquired by the antenna, the received signal is typically input to a receiver front-end circuit, integrated circuit or module. The receiver front-end circuit typically comprises a Low-noise Amplifier (LNA) followed by down-conversion circuitry and associated filtering. Often, the LNA is located after an antenna switch or duplexer in a transceiver arrangement, whereby the antenna switch or duplexer separates transmit signals from receive signals passing through the antenna. The purpose of the LNA is to amplify, as much and as linearly as possible, the desired signal that is captured by the antenna. The LNA not only deals with weak desired signal but also large interferences (such as transmitter leakage attenuated not enough by the duplexer). Using an LNA, the effect of noise from subsequent stages of the receiver chain is reduced by the gain of the LNA, whilst the noise from the LNA device/circuit itself is injected directly into the received signal. Thus, noise created within the LNA is a critical factor in its design and, hence, the received signal is required to be amplified without adding noise or distortion and without the LNA consuming too much power. In this way, retrieval of the received signal is possible in the later (baseband, following down-conversion) stages of the receiver.
A good performance LNA usually features a current-mode output interface (acting as a transconductance amplifier (sometimes referred to as a ‘transconductor’)) and has a low-noise factor (‘NF’, sometimes referred to as ‘F’) (typically of the order of 1 dB), and should have large enough intermodulation and compression point (third order intercept point (IP3) and 1 dB compression point (P1dB)) performance. Further important criteria in the design of an LNA includes: operating bandwidth, gain flatness, stability and input and output voltage standing wave ratio (VSWR). A further critical factor in LNA design is to obtain a good impedance match between the antenna and the LNA input. Often, the solution to this potential problem has been to use on-chip LNAs using a common-source coupled transistor with inductive source degeneration. The inductor at the source resonates at the desired frequency to obtain real valued input impedances. However, due to the resonance effect of inductor circuits, the bandwidth for this type of LNA is low and sometimes unacceptable for cellular phone application that must be able to receive signals over a wide range of frequencies, say from 850 MHz to 2.5 GHz.
In current cellular phone handsets, one solution is to implement a number of LNAs in order to cover the whole of the desired bandwidth. A switch activates the appropriate LNA according to the frequency to be received. In this solution each LNA has its own inductors, which provides good gain and low-noise, but at the cost of a larger integrated circuit. Inductors use a lot of area on chip and all the matching components off chip use valuable PCB area. To make matters worse inductors on chip require expensive manufacturing steps in order to achieve a high ‘Q’ factor.
Hence, there has been a recent trend to investigate receiver front-end designs that are inductor-less, which often employ noise cancellation in order to achieve low-noise. However, it is known that in designing LNAs without inductors (i.e. inductor-less), it is typically very hard to achieve high linearity together with a good noise factor (NF), since the ‘active’ input matching that is required, due to the removal of inductors, in order to provide efficient signal power transfer, can adversely limit both the linearity and noise figure (NF) performance of the LNA.
FIG. 1 illustrates a simplified circuit diagram of a receiver front end circuit 100 comprising a noise-cancelling cascode CMOS LNA arrangement, as described in U.S. Pat. No. 8,503,967 B2. The primary purpose of the noise-cancelling function in the illustrated receiver front end circuit 100 is to achieve an inductor-less front-end design. Here, to illustrate the NF limiting effect of an inductor-less LNA, the receiver front end circuit 100 comprises a source (represented by an input signal 105 with a source resistance (RS) 110). A cascade arrangement for CMOS transistors 115, 120 comprises a feedback resistance RFB 125. The receiver front end output 140 is illustrated as being a summation of the output of first and second transconductance amplifiers 130, 135. With the input signal matched to the source resistance (RS 110), it is known that the noise factor F of the receiver front end circuit may be derived as follows:
Input matching of source resistance (RS 110) may be defined as:
                              R          S                =                                            R              FB                        +                          r              oa                                            1            +                                          g                ma                            ⁢                              r                oa                                                                        [        1        ]            Where:
RS=Source resistance;
RFD=Feedback resistance;
rOA=the internal resistance of the transconductance operational amplifiers; and
gma=gain of the CMOS transistors 115, 120 (Ma1/Ma2).
In order to cancel noise generated in CMOS transistors 115, 120 (Ma1/Ma2), the gain (Gm2) of the second transconductance amplifier 130 may be defined as:
                              G                      m            ⁢                                                  ⁢            2                          =                                            R              S                                                      R                FB                            +                              R                S                                              ⁢                      G                          m              ⁢                                                          ⁢              1                                                          [        2        ]            
Thus, it can be readily appreciated that the gain of the second (noise cancelling) transconductance amplifier 130 is controllably less than the gain of the main (first) low-noise transconductance amplifier 135. The noise figure of the receiver front end circuit may then be defined as:
                    F        =                              1            +                                                                                                                                                                                    4                            ⁢                                                                                                                  ⁢                            γ                                                                                                              G                                                              m                                ⁢                                                                                                                                  ⁢                                1                                                                                      ⁢                                                          R                              S                                                                                                      ⁢                                                  (                                                                                    1                              +                                                                                                R                                  FB                                                                /                                                                  R                                  S                                                                                                                                                    1                              +                                                                                                (                                                                                                            2                                      ⁢                                                                                                                                                          ⁢                                                                              R                                        FB                                                                                                              -                                                                          1                                      /                                                                              g                                        ma                                                                                                                                              )                                                                /                                                                  R                                  S                                                                                                                                                                                        _                                        )                                                                              G                                              m                        ⁢                                                                                                  ⁢                        1                                                              ⁢                    noise                                                  2                            ⁢                              (                                  1                  +                                      1                                                                                            1                          +                                                                                    R                              FB                                                        /                                                          R                              S                                                                                                      _                                                                                              G                                                      m                            ⁢                                                                                                                  ⁢                            2                                                                          ⁢                        noise                                                                                            )                                      +                                          4                ⁢                                                                  ⁢                                  R                  FB                                ⁢                                  R                  S                                                            (                                                                                                                                                          2                            ⁢                                                                                                                  ⁢                                                          R                              FB                                                                                +                                                      R                            S                                                    -                                                      1                            /                                                          g                              ma                                                                                                      )                                            _                                                                                      R                        FB                                            ⁢                      noise                                                        2                                                              ❘                                    [        3        ]            Where:
γ=bias-dependent thermal noise factor;
From equation [3], it is clear that the noise of the devices used for input matching (Ma1 and Ma2) is cancelled and the noise figure is now dominated by the noise in the main (first) low-noise transconductance amplifier 135.
FIG. 2 illustrates a simplified representation 200 of a noise-cancelling cascode CMOS receiver front end circuit, highlighting a linearity bottleneck effect of an inductor-less LNA problem, as identified by the inventors. In effect, a linearity bottleneck may occur in the circuit's operation if a third transconductor (Gm3) is an out-of-phase gm cell, compared with the main (first) transconductor Gm1 135. In one example, the 3rd transconductor may not be low-noise as its noise is suppressed by the voltage gain amplifier 230, whereas the 1st (main) transconductor may be low-noise.
if we assume that gmaroa>>1 and RFB>>roa, the noise factor F of the circuit of FIG. 2 may be defined as:
                    F        ≅                  1          +                      γ                                          G                                  m                  ⁢                                                                          ⁢                  1                                            ⁢                              R                S                                              +                                    R              S                                      R              FB                                                          [        4        ]            
The two basic problems of the 1st-generation noise-cancelling receiver front end circuit are:
(i) Low intrinsic gain of the input-matching voltage gain amplifier 230, which harms the noise generated from RFB 120, low-noise transconductance amplifier Gm1 135, and noise-cancelling transconductor Gm3 205. Notably, this situation worsens for higher radio frequencies and in more advanced CMOS processes.
(ii) The voltage gain of voltage gain amplifier 230 is necessary for the shunt-shunt feedback used to provide input 50 ohm matching. It therefore may produce a large voltage swing at the input of the third transconductor (Gm3), 205, thereby degrading the linearity exhibited by the third transconductor (Gm3) and causing a potential linearity bottleneck in the circuit (e.g. making the circuit operate nonlinearly):
Referring now to FIG. 3, a further simplified known LNA circuit 300, employing a frequency-translation feedback receiver is illustrated. The LNA circuit 300 comprises a radio frequency input 305 to a transconductance amplifier (Gm), 310. In contrast to the architecture shown in FIG. 1, where the shunt-shunt feedback gain required for input matching is provided by a single-stage transconductor (Ma1 and Ma2), the gain used for feedback in this figure is provided by the entire receiver front-end. Therefore, the feedback resistor Rf would typically be quite large to reduce its noise. The noise figure can be proved to be the same as that described in [4]. One problem of the architecture of FIG. 3 is the nonlinearity generated by the TIA, due to the large voltage gain (and hence swing) at the TIA output.
Thus, a need exists for an improved higher-performance inductor-less LNA, for example suitable for next-generation receivers, and a method of operation therefor. In particular, a need exists for an improved receiver front-end architecture that exhibits better performance and/or dynamic range with respect to NF and linearity.